C H A P T E R 4

Microwave Link Engineering

Microwave link engineering is the branch of communication engineering that deals

with the analysis, design, implementation, and testing of radio paths in the atmosphere

or in space, operating in the range of about 1 to 100 GHz. Satellite and

other microwave radio links obey certain rather predictable laws of nature, allowing

engineers to design satellite networks on paper (or, more likely, on computer) with

good accuracy. An unobstructed line-of-sight path provides the most predictable

transmission link; however, various forms of signal absorption (attenuation),

blockage, and multipath propagation (caused by reflections) introduce time-varying

fading, which can best be estimated using statistical techniques. Nevertheless, the

service obtained from microwave links between the satellites and ground stations

can be very acceptable, even when compared to terrestrial cable and radio (wireless)

systems.

This chapter is intended for engineers and scientists new to the field and is

perhaps the most mathematical chapter in the book. Nontechnical readers may

wish to scan it rather than delve into the details. In general, microwave energy

travels in straight lines through space in the same manner as a light beam. That

should not be a surprise because both light and microwaves are forms of electromagnetic

energy, light being at much higher frequencies or, equivalently, shorter wavelengths.

Radio waves in general and microwaves in particular can propagate through

space like light and heat, spreading out as they move farther and farther from the

source. The waves travel through a vacuum at the universal speed of light, the

same for all forms of electromagnetic energy. Microwaves can be kept from propagating

freely by forcing them to travel in a closed metallic pipe called a waveguide.

The physical similarity to common household pipes has led to the use of the term

plumbing when referring to the waveguides found in Earth stations and satellites.

At the input end of the waveguide would be found the transmitter, the electronic

device that generates the microwave energy on the proper frequency, with the other

end connected to an antenna such as a parabolic ‘‘dish.’’ Waveguides are similarly

used on the receiving end.

4.1 The Decibel

Microwave power is measured in watts, the same unit we associate with electric

power used for lighting and motors. Link effects that either reduce or increase that

power almost always are expressed in terms of the decibel (dB), which is why it

109

110 Microwave Link Engineering

is important to introduce the concept early in the discussion. Converting a quantity

to decibels simply means taking the logarithm to the base 10, or common logarithm,

and multiplying the result by 10:

A (in dB) = 10 log10 (a)

where a is a real, nonnegative number. In actuality, a is a ratio, such as the power

gain of an amplifier, that is,

a = Pout /Pin

A sample of decibel values versus their equivalent ratios is provided in Table

4.1. Recall that adding logarithms is equivalent to multiplying ratios. So, for

example, adding 4.0 dB is equivalent to multiplying by a factor of 2.51. In the

table, relative power changes are computed to the closest tenth of a decibel (i.e.,

0.1-dB precision) because measurement accuracy on an actual satellite link is about

0.5 dB at best. Since the logarithm of the number 1 is equal to zero, a zero-dB

change simply means there has been no change in the power level of the signal.

You may recall that logarithms are exponents, and in the case of logarithms to

Table 4.1 Example of a Decibel Table,

Indicating the Ratio, Percentage

Increase, and Corresponding Decibel

Value

Power Ratio % Increase dB

1.0 0 0

1.12 12 0.5

1.26 26 1.0

1.41 41 1.5

1.59 58 2.0

1.78 78 2.5

2.00 100 3.0

2.24 124 3.5

2.51 151 4.0

2.82 182 4.5

3.16 216 5.0

3.55 255 5.5

3.98 298 6.0

4.47 347 6.5

5.01 401 7.0

5.62 462 7.5

6.31 531 8.0

7.08 608 8.5

7.94 694 9.0

8.91 791 9.5

10.0 900 10.0

20 13.0

50 17.0

100 20.0

200 23.0

500 27.0

1000 30.0

4.2 Propagation on the Earth-Space Link 111

the base 10, they are exponents of the number 10. The inverse of a decibel is

expressed as:

a = 10A/10

which again is a ratio.

A decrease would be represented by a negative decibel value. For example,

reducing a power to one-half its value (e.g., multiplying it by 0.5) is the same as

subtracting 3 dB (e.g., adding 3.0 dB). Another consequence is that a decrease of

1.0 dB (adding 1.0 dB) represents a decrease of 20%. Table 4.1 indicates that

adding 1.0 dB is an increase of 26%.

An inexpensive scientific calculator is the most convenient device for converting

back and forth between ratios of powers and decibels. Just be sure to use the

common logarithm (log) button and the 10X button, where appropriate. This is

safer than trying to memorize Table 4.1 (although such knowledge is the sign of

a ‘‘dB artist’’).

The following are a few simple examples of the decibel approach. A 3-dB

difference in power level is nearly equal to a factor of 2, while a factor of 10 is

exactly equal to 10 dB, that is, 10 log (10) = 10. In satellite communications, it is

not uncommon to encounter a factor of 1020 decrease in signal power. That can

more conveniently be expressed in decibels as a 200-dB reduction, called a loss.

Gains are added, while losses are substracted. When dealing with factors in an

equation, it is convenient to first convert each factor to decibels. Then the equation

can be solved by summing the decibel values instead of having to resort to multiplication.

Likewise, division becomes simple subtraction, as in the example of the

200-dB loss. That is the basis for the link power balance calculation and the link

budget, reviewed later in this chapter.

4.2 Propagation on the Earth-Space Link

The process by which radio signals reach the receiving antenna from the transmitting

station is called radio wave propagation. Radio waves represent a part of the

electromagnetic spectrum, ecompassing radio, infrared, visible light, ultraviolet, and

X rays (given in increasing order of frequency or decreasing order of wavelength).

The part of the electromagnetic spectrum that is of interest in commercial

satellite communications lies between 1 and 60 GHz, as illustrated in Figure 4.1.

Figure 4.1 The microwave spectrum, indicating the approximate location of satellite bands.

112 Microwave Link Engineering

The relative merits of each band segment were covered in Chapter 1. Using the

popular letter designations for the various segments, the spectrum offers a total

bandwidth of approximately 60 GHz. The frequencies below 30 GHz are those

currently in use or under development; the frequencies above 30 GHz are a still

untapped resource. RF bandwidth can be used and reused many times by multiple

satellites and the radio beams that they can generate over portions of the Earth. The

following sections review the propagation, generation, and reception of microwave

signals, in preparation for studying the unique aspects of satellite communications

presented in Chapters 5, 6, 7, and 9.

4.2.1 Basic Microwave Propagation

Once it is several emitter dimensions away from its emitter, electromagnetic wave

energy propagates out radially. The spreading pattern decreases in intensity

inversely to the square of the distance. All radio waves behave in this manner in

free space, but various forms of matter produce interesting and potentially disruptive

results when placed in their path. That is because microwave energy can be

absorbed, scattered, bent, and reflected, as indicated in Table 4.2. As a result, there

is additional loss of signal power, in decibels, which must be accounted for in the

link design.

A question that comes to mind is, ‘‘How does such a wave, traveling at the

speed of light, come into existence in the first place?’’ Because space consists of

nothing, a radio wave does not propagate like the crest of a wave in the ocean.

Instead, it usually is the result of the high-frequency vibration of electrons in a

piece of wire or other conducting material of appropriate dimensions. Alternating

current from a transmitter causes the electrons to vibrate back and forth; by not

flowing continuously in one direction, the electrons lose most of their energy by

throwing it off into space. That is illustrated in the drawing sequence provided in

Figure 4.2.

A high transmitter power, say, 1,000W (1 kW), produces a correspondingly

high level of microwave radiation from the antenna at the point of exit into space.

That is the same principle behind a microwave oven, which uses a high-power

Table 4.2 Propagation Effects to Introduce Additional Attenuation of Microwave Signals as They

Propagate Between the Transmitting Station and Receiving Station

Propagation Mode Effect Source Medium

Absorption Conversion of microwave energy Air, water vapor, precipitation

into simple heat (rain and snow), nonmetallic

structures, and foliage

Scattering Multiple reflections and bending Water vapor, precipitation, dust,

due to minute particles ion plasma (the ionosphere)

Refraction Bending of microwaves due a Air, ion plasma

change in the refractive index

Reflection Total reflection of the microwave Temperature inversion layers

signal (ducting), metal structures, bodies

of water, buildings, and flat terrain

Diffraction Bending around physical obstacles Metal structures, buildings, and

partial terrain blockage

4.2 Propagation on the Earth-Space Link 113

Figure 4.2 The fundamental concept of electromagnetic wave propagation from an elemental

antenna. The packages of energy are separated by one wavelength in distance.

magnetron amplifier to generate an intense field for cooking purposes. The field

causes the molecules in the food to oscillate at the same frequency; that generates

the necessary heat to raise the temperature. Such intense radiation around a highpower

microwave transmitting station can ‘‘bake’’ you if you come too close. At

sufficient distance from a transmitting antenna, the microwave energy induces less

heating. There is a defined area around the antenna, called the near-field region,

where the energy has not quite coalesced into a clearly defined beam. Within the

near field, microwave energy varies in intensity, depending where one stands relative

to the antenna structure. At sufficient distance from the antenna, the radiation

field is formed into the type of beam associated with the particular antenna structure,

called the far-field region. A parabolic reflector antenna has a sharply defined

far field beam that is aimed at the satellite. The transition between the two regions

is roughly defined by the following equation:

Rf 2D2/_

where Rf is the approximate distance, D is the diameter of dish reflector in meters,

and _ is the wavelength in meters. This equation indicates something unexpected:

the distance to the transition increases with frequency rather than decreasing.

For example, proper testing of a Ku-band reflector antenna demands that the

transmitting (or receiving) source be farther away than if the same reflector were

to be used at C-band. From the beginning of the far field, the radiated signal keeps

on decreasing in intensity by the inverse of the square of the distance from the

source (the same property is visible light from a point source such as a star). This

particular concept is the basis for accurate link design.

4.2.2 Isotropic Radiator

The most fundamental type of radio antenna is the isotropic source, which is

analogous to the lightbulb illustrated in Figure 4.3. At a fixed radius from an

114 Microwave Link Engineering

Figure 4.3 An isotropic source can be represented by a lightbulb radiating energy in all directions

with equal intensity.

isotropic source, which defines a sphere, the energy intensity is constant regardless

of direction. The area of the sphere of uniform received energy is equal to the

constant _ (3.14159...) multipled by 4 multiplied by the square of the radius, that

is, A = 4_R2. It is common practice to measure the signal intensity at the particular

radius in units of watts per square meter, calculated by dividing the power of the

isotropic source by the area of the sphere in square meters:

P/A = Pt /4_R2, W/m2

The denominator, 4_R2, is sometimes called the spreading factor.

RF power driving the isotropic source produces a constant power density at a

fixed distance. That density decreases by 1/R2 as the point of reception moves

farther away from the source. Ignoring losses, it is theoretically possible to capture

all the transmitted power by collecting it with a closed surface around the source

(the physical size of the collector is immaterial as long as it closes on itself).

Receiving antennae, therefore, work by way of the area that they expose to

the RF energy radiating from the source. This is illustrated in Figure 4.4 by an

isotropic source that radiates energy equally in all directions, shown as equally

spaced rays. There are two receiving antenna surfaces of equal area: one at distance

R2 is farther away from the source than the other at distance R1. Notice how the

closer antenna intercepts considerably more power than the more distant area.

That illustrates how a radio signal becomes weaker as the receiver is moved farther

from the transmitter. It also demonstrates the important concept of capture area,

that is, the relationship between the effective area of an antenna and the strength

of the signal received by it. The challenge of antenna design is to maximize the

fraction of the energy that the antenna actually delivers from the reflector surface

to the receiver. The key parameter is the efficiency of the antenna, defined as the

ratio of the effective area (i.e., the area that would perfectly capture the same

amount of energy) to its physical area. Typical values of dish antenna efficiency

4.2 Propagation on the Earth-Space Link 115

Figure 4.4 The radio energy captured by a fixed antenna area decreases as the distance to the

source increases.

are between 55% and 70%. Antennas that do not use reflectors but employ an

active receiving area to the incoming wave can achieve up to 90% efficiency.

Examples include horns and arrays of elemental antennas.

4.2.3 Directional Properties of Antennas

The isotropic antenna is neither practical (its ideal characteristic cannot be achieved

with a simple physical structure) nor particularly useful for satellites (due to its

low gain and inability to discriminate directions). What we rely on for receiving

and transmitting signals is the concept of directivity, which simply means that the

antenna has the ability to focus the energy in specific directions. Energy that would

have been radiated in unnecessary directions around the sphere is concentrated by

the structure of the antenna and redirected to increase the intensity in the desired

direction.

4.2.3.1 Directivity and Gain

The directivity in a given direction can be expressed as the ratio of the measured

signal to the maximum signal in the peak direction. The gain, on the other hand,

is an absolute measure, obtained by comparing the signal from the antenna to that

of an isotropic radiator. Our absolute measure of maximum gain could be as little

as 10 or greater than a factor of 1 million. When expressed in decibels, these

particular values would be 10 dBi and 60 dBi, respectively, where the ‘‘i’’ indicates

that the gain is with respect to our isotropic antenna. That implies that an isotropic

antenna has a gain of 0 dBi. There must be a fixed offset in decibels between the

absolute value of gain and the corresponding relative value of directivity, because

our definition of directivity is that it starts at a value of 0 dB at the peak and

follows the pattern in a negative-going sense for all other angles.

Another important property of an antenna is called reciprocity, which means

that the gain and the directivity are the same at a given frequency whether it is

used to receive or to transmit. That allows the antenna to receive with precisely

the same directional characteristics as it transmits. Figure 4.5 illustrates a simplified

antenna gain pattern for transmission or reception with a main beam (the region

116 Microwave Link Engineering

Figure 4.5 Gain of a high-gain antenna expressed as a power ratio and presented as a 360-degree

polar plot.

of maximum directivity) oriented toward the right. For comparison, the uniform

pattern of an isotropic source is superimposed to scale on the directional antenna

pattern. In addition to a main beam, every real antenna operates in undesired

directions, shown in the figure as a pair of sidelobes and a backlobe. The maximum

gain, also called the peak gain, is indicated at the center, and the backward direction

is indicated at }180 degrees.

4.2.3.2 Beamwidth and Sidelobes

There are a number of other useful definitions of antenna performance besides the

peak gain. The half-power beamwidth (often called simply the beamwidth) is the

angular width of the main beam measured between the points where the power

intensity is one-half that of the peak. An equally accurate name that is often used

is the 3-dB beamwidth, since the half power point is where the directivity is 3 dB

down. Assuming that the microwave link can still function with a 3-dB decrease

in signal strength, the half-power beamwidth defines the range of antenna pointing

(alignment angle) over which the antenna or satellite can move. It is a common

practice, however, to allow only a 0.5-dB (12%) drop in signal power, which

demands either tighter antenna pointing accuracy or satellite position control.

When installed at an Earth station, the antenna is attached to a mount, which

may allow the beam to be repointed. Simple RO antennas used for direct reception

usually are fixed to their mounts and need only be pointed toward the satellite

when initially installed. Movable antennas usually have motorized mounts to allow

remote repointing; that could also be part of an automatic tracking system to

initially locate (acquire) a satellite and to keep the beam aligned with (track) the

satellite. It is normal to refer to the elevation angle of the direction of pointing,

where zero is for pointing at the horizon and 90 degrees corresponds to directly

overhead (zenith). The mount also points the antenna in azimuth, that is, along

the directions of the compass. Most mounts work directly in terms of elevation

and azimuth, but some are polar and behave like an astronomer’s telescope. The

advantage of a polar mount is that it can repoint along the equatorial orbit plane

by adjusting only one axis.

Presentations of antenna performance are called antenna patterns and are

illustrated in Figure 4.6 (a hypothetical example) and Figure 4.7 (a real 10-m

antenna) in terms of gain and directivity, respectively. Almost every antenna has

a backlobe in the opposite direction from the main beam. However, as shown in

Figure 4.6, the gain of the backlobe can be made to be less than unity, in this case,

producing a negative gain of 2 dBi or lower. Sidelobes and backlogs are important

4.2 Propagation on the Earth-Space Link 117

Figure 4.6 Gain of a directional antenna in decibels for all angles with respect to the peak of the

main beam indicating the worst-case envelope of sidelobes.

Figure 4.7 A comparison of the radiation (directivity) pattern of a real 10m C-band antenna with

the ITU sidelobe envelope.

characteristics of Earth station antennas when one is considering transmit or receive

interference. Figure 4.7 includes a smooth curve, called a sidelobe envelope, that

defines a specification of maximum sidelobe gain. Two such envelopes are shown,

the lower being a tighter specification that applies between 1° and 20° off the peak

of the main beam.

4.2.3.3 Isolation

The directive property of an antenna dertermines how effective it will be for getting

signal power from the source to the receiver. However, any link can be degraded

by signals on the same frequency that enter the receiving antenna from a direction

118 Microwave Link Engineering

other than along the main beam. Likewise, a transmitting station can cause degradation

to other systems by sidelobe radiation. Any undesired signal that can potentially

degrade reception is RFI. There is a whole field of engineering study that focuses

on the identification of sources of RFI, the establishment of criteria for acceptable

operation in the presence of RFI, and the development of techniques for countering

its effects. Techniques such as beam shaping, cancelation, and shielding are effective

in that regard.

The presence of RFI is a consequence of the fact that all radio communications

systems reuse frequencies, that is, there will be more than one radio station operating

on any particular frequency at any particular time. What keeps the RFI within

acceptable limits is isolation that is either natural (e.g., geographical or angular/

orbital separation) or artificial (e.g., measures such as RFI shielding or beam

cancellation). Satisfactory operation of independent microwave users on the same

frequency often requires cooperation (called frequency coordination) because ‘‘one

user’s radio link is another user’s RFI.’’ Several satellites can operate in the geostationary

arc in the same frequency band because each directional ground antenna

can focus on one particular satellite, suppressing the RFI produced by adjacent

satellites (see Figure 1.26).

For example, if we assume the receiving user’s antenna has the characteristic

given in Figure 4.6 and the interfering satellite is located 45 degrees away from

the satellite that we wish to receive (the desired satellite), then the RFI is suppressed

by at least 20 dB. Actual ground antennas used in FSS and BSS links provide 40

to 60 dB of peak gain with suppression at 45 degrees of 50 to 70 dB. At closerin

angles, the isolation may amount to only 20 to 30 dB, which is adequate to

allow satellite spacing as small as 2 degrees. The first sidelobe, in particular, is

normally the strongest, being typically 15 to 20 dB down from and within 1 degree

(more or less) of the peak gain. The control of those sidelobes demands careful

design and installation of such antennas. A useful specification in that regard is

the sidelobe envelope (shown in Figure 4.7), which defines the worstcase potential

for RFI. A standardized formula, which was adopted by the ITU as Recommendation

S.465-5, provides a common standard for the larger ground antennas used at

C and Ku bands. In it, gain at a particular off-axis angle is specified in the direction

of a potentially interfering (or interfered-with) satellite by:

G(_) 29 25 log10 _, dBi

where _ is the offset angle between the direction of the main beam and that toward

the interfering or interfered-with satellite, for angles between }1 degree and

}37 degrees. Furthermore:

G(_) 8 dBi, for angles between }7 degrees and }9.2 degrees

For angles between }9.2 degrees and 48 degrees:

G(_) 32 25 log10 (_), dBi

Lastly, G(_) ≤ −10 dBi for _ > 48°.

4.2 Propagation on the Earth-Space Link 119

4.2.4 Polarization (Linear and Circular)

Up to this point, we have dealt with radio propagation as if it were pure energy.

However, there is a property of an electromagnetic wave called polarization, that

depends on the orientation (or angle of rotation) of the transmitting antenna. You

may be familiar with polarization as it relates to light. For example, true threedimensional

(3-D) movies utilize vertically and horizontally polarized light to simultaneously

project appropriate left and right images on the screen. The images are

separated by invisible polarizing grids imbedded in the lenses of the viewer’s glasses,

where one lens is aligned vertically while the other is aligned at a 90-degree angle

(horizontally). Vertical and horizontal polarization, therefore, can ‘‘reuse’’ (i.e.,

use twice) a transmission path, such as the projection of a movie on a screen or,

as described below, a radio path.

4.2.4.1 Linear Polarization

The concept of polarization discrimation in radio communications is illustrated in

Figure 4.8. Shown at the top of the figure is a type of simple wire or rod antenna

called a dipole. Electrical current from the transmitter flows along the rod first

upward and then downward, oscillating at the frequency of transmission. At C-band

downlink frequencies, the rate of oscillation is 4 billion times per second (i.e.,

4 GHz). The alternating current in the rods produces an electromagnetic wave that

propagates off into space (Figure 4.2). A dipole is not a true isotropic source

because there is no radiation in the direction the rods point. Instead, what is formed

is a doughnut-shaped pattern that is aligned horizontally. The electrical currents

in the rods cause the electromagnetic wave to have its electric component to be

lined up in the same direction, which is vertical (in the direction of the two arrows).

This type of polarization is called linear polarization (LP) because the electric

component has a fixed orientation. Horizontal LP is obtained when the dipole is

rotated 90 degrees, so that the direction of the electrical current also is horizontal.

Reception occurs when the electric component of the incoming wave produces a

Figure 4.8 Properties of linear polarization as radiated and received by dipole rod antennas.

120 Microwave Link Engineering

current in the receiving antenna, which cannot occur if the conductors of the

receiving antenna are perpendicular with the incoming polarization. In the lower

half of Figure 4.8, horizontally polarized transmitting and receiving antennas provide

for a maximum amount of power to be carried (coupled) between them. A

vertically polarized receive antenna, which is perpendicular to and therefore crosspolarized

with the transmitter, minimizes the amount of coupled energy.

Microwave antennas usually make use of waveguide structures and solid

reflecting surfaces because they are much more efficient and predictable than wires

and rods. For example, if one encloses the dipole antenna in a metal box, it can

feed a waveguide. Rectangular waveguide, illustrated behind the horn in Figure

4.9, is simple in design and efficient in the transmission of microwave energy and

hence is very popular in satellite and Earth station design. The electric component

is vertical for the orientation shown, extending between the upper and lower side

walls (dimension b). The magnetic component lies in the plane of the top and

bottom walls and is perpendicular to the electric component at every point along

the waveguide. The horizontal dimension of these walls (dimension a) must be

greater than one-half wavelength at the operating frequency. At any lower frequency,

the wave cannot propagate because its wavelength literally cannot fit. This

is termed waveguide beyond cutoff. The narrower dimension, b, is not as critical

and is adjusted like a transformer to match the characteristic impedance of the

waveguide to the antenna.

The relationship between horn antenna orientation and wave polarization is

shown in Figure 4.9, which basically is a slightly flared piece of rectangular waveguide.

Instead of electrical currents, the waveguide carries a bound electromagnetic

wave with the electric components extending between the parallel walls. Creation

of the guided waves in the first place is accomplished with the small monopole

element inserted into the waveguide at the input end. Note how the electric components

vary in intensity along the width of the horn, being maximum at the center

(closely spaced arrows) and dropping to zero at the edges. That particular natural

Figure 4.9 Linear polarization of microwave radiation from vertically and horizontally polarized

feed horns.

4.2 Propagation on the Earth-Space Link 121

property of a simple horn provides a radiation taper used in the design of hornfed

reflector antennas. In any case, the radiated electromagnetic wave is linearly

polarized, just like that from a vertically oriented dipole. The polarization can be

changed to horizontal simply by rotating the horn and waveguide by 90 degrees.

As with wire antennas, maximum coupling occurs when the transmitter and

receiving horns are co-polarized. If one horn is rotated 90 degrees with respect to

the other, minimum energy transfer (coupling) results and they are said to be crosspolarized.

The performance of LP at intermediate points follows a simple law: The

relative energy coupled is equal to the square of the cosine of the angle. That

characteristic is plotted at the left of Figure 4.10 for the copolarized case, that is,

for the level of signal received as the receiving antenna is rotated from maximum

coupling to minimum coupling. An ideal LP wave and antenna are assumed, for

which the coupling goes from a maximum of 1 to a minimum of zero. Maximum

coupling always occurs at some angle, and minimum coupling is as close to zero as

physically possible. Properly designed and installed antennas can deliver minimum

coupling values of 0.0001, or 40 dB. Larger structures that are highly symmetrical

in the direction of radiation can even exceed that value. However, due to slight

misalignments and other imperfections, values in the range of 30 to 40 dB are

more typical.

Probably the most important application of polarization is in frequency reuse,

where two cross-polarized signals are transmitted at the same time on the same

frequency. The right curve in Figure 4.10 shows how the level of the cross-polarized

signal increases as the receiving antenna is rotated from 0 to 90 degrees. Notice

how at 45 degrees both signals are at the same level, which results in no isolation

between the signals on opposite polarizations. Figure 4.11 plots coupling in decibels,

termed polarization isolation, between the desired and undesired polarizations.

Maximum isolation occurs at zero offset angle, that is, where the receiving antenna

is aligned in polarization with the transmitting antenna and the undesired polarization

is nulled out (minimized). Alignment for maximum coupling is not particularly

Figure 4.10 Relative receive power as a true ratio for the copolarized and cross-polarized signals

in a linearly polarized link as the polarization angle of the receiving antenna is rotated.

122 Microwave Link Engineering

Figure 4.11 Polarization isolation in decibels for a linearly polarized RF signal as a function of the

offset angle from maximum isolation.

critical (}5 degrees of error introduces very little loss of signal); however, the crosspolarization

isolation is extremely critical, and precise alignment is a necessity. The

maximum value of isolation of 40 dB in Figure 4.11 corresponds to the typical

value described in the preceding paragraph. All FSS and BSS satellite communications

systems use polarization isolation to increase the capacity of the particular

orbit position. Therefore, users must employ antennas on the ground that meet

polarization requirements; linear polarizations should be adjusted from time to

time by rotating the feedhorn for minimum RFI. (The only way around the latter

requirement is to use circular polarization, discussed in the next section.) Isolation

can subsequently degrade from propagation effects, particularly Faraday rotation,

which is a twist of linear polarization caused by the ionosphere. If the ionosphere

caused a change of 6°, then the cross-polarization isolation would be approximately

20 dB. Such a low value generally is considered to be unacceptable.

4.2.4.2 Circular Polarization

Another type of polarization that is used in international, DTH broadcasting and

MSS satellites is circular polarization (CP). Being circular in polarization, the

receiving antenna is not aligned for the transmitting antenna’s polarization. Rather,

it is the inherent polarization of the signal, either right-hand CP (RCP) or lefthand

CP (LCP), that must be supported by both ends of the link. An example of

RCP is provided in Figure 4.12.

To understand the composition of CP, first observe that LP can be represented

in the form of a vector, where the direction of the vector is in line with the electric

component and its magnitude is proportional to the square root of the power of

the signal. This vector is always perpendicular to the direction of propagation. CP

is a particular combination of two equal LP waves that are cross-polarized with

respect to each other (horizontal and vertical are two of the four vector states in

Figure 4.12). If the two polarizations represent the same signal on the same frequency

and they are precisely in phase with each other, the resultant vector lies at

4.2 Propagation on the Earth-Space Link 123

Figure 4.12 Rotation of the electric component during the propagation of a circularly polarized

RF signal.

a 45-degree angle between the two polarizations with an amplitude that is 2

times either amplitude. In CP, however, the same two vectors are out of phase

with each other by 90 degrees. That is accomplished by first splitting the transmit

signal in two at the source antenna and then delaying one component by a quarter

period before radiating the components through a dual-polarized antenna element.

What is produced is a resultant vector that rotates like a corkscrew (Figure 4.12)

as it propagates through space. The sense of polarization is either rotating to the

right (RCP) or to the left (LCP), looking in the direction of propagation. That is

determined by which linear component is delayed with respect to the other. Frequency

reuse is possible because LCP and RCP waves are oppositely polarized with

respect to each other, just like vertical and horizontal LP waves.

The advantage of CP is that the receiving antenna does not need to be aligned

in polarization since the electric component rotates 360 degrees at a rate equal to

the frequency of transmission. A receiving antenna could consist of combined

vertical and horizontal LP elements, allowing the components to be coupled and

recombined with the proper phase relationship. The sense of combining determines

whether RCP or LCP will be recovered. Because most waveguides are linearly

polarized, a device called a polarizer is used to make the conversion to CP. Also,

the waveguide must be square or round to prevent cutoff at the operating microwave

frequency. The most common approach is to generate two orthogonal linear polarizations,

delay one of them by 90 degrees (one-quarter wavelength), and then add

them. In general, the amount of polarization isolation thus obtained is not as high

as that achieved by LP because of the imbalance between the two components that

often results in practice. This imbalance produces an elliptical pattern measured

in terms of the axial ratio. (CP has an axial ratio of 1, or 0 dB.)

4.2.5 Propagation Losses

The free-space path between transmitter and receiver may pass through a combination

of ground obstructions and atmospheric phenomena. They can be divided into

124 Microwave Link Engineering

those that are constant (and usually predictable) and those that are both time

variant and random (which may be estimated statistically). Taken together, these

losses reduce the strength of the received signal and can cause its level to vary over

time even to the point where the signal fades away. This is illustrated in Figure

4.13, which shows received signal power (signal level) as a function of time during

a period of rapid variation (fading). The mean signal level results from the constant

and predictable losses on the link; the random losses cause the level to decrease

and occasionally drop below the acceptable threshold. When the signal dips below

that threshold, it could become unusable for communication purposes because of

excessive noise and, in the case of digital links, an undesirable high error rate and

ultimately a loss of synchronization as well. The difference in decibels between the

mean level and the threshold is called the link margin, which is discussed at the

end of this chapter.

Table 4.3 summarizes the primary losses that must be determined and accounted

for in any microwave link design. The table compares typical downlink values for

L-, C-, Ku-, and Ka-bands. Certain losses are critical in all satellite systems, while

others are of concern only in specific cases.

4.2.5.1 Free-Space Loss

We already have covered in some detail the mechanism for free-space loss, which

diminishes signal strength due to distance alone. For 1 m2 of effective antenna

area, the captured energy is simply

Pr = Pt /4_R2

where Pt = the radiated power in watts and R = the distance in meters.

The typical distance from a point on the Earth to a satellite in GEO is 36,000

km (22,300 miles).

This relationship, which should be recognized from the previous discussion of

isotropic radiation, does not depend on frequency. It is a common practice to

introduce frequency into the equation by splitting apart the propagation loss from

the gain of the receiving antenna (here assumed to have an effective area of 1m2).

Figure 4.13 Variation of the received carrier power on a microwave link during a period of severe

fluctuation (fading).

4.2 Propagation on the Earth-Space Link 125

Table 4.3 Comparison of Significant Propagation and Fading Modes and Resulting Losses That Affect

Microwave Links over GEO Paths (Values Are Rough Order of Magnitude, in Decibels)

Propagation Relative Importance L-Band C-Band Ku-Band Ka-Band

Mode on Satellite Links (1.6/1.5 GHz) (6/4 GHz) (14/12 GHz) (30/20 GHz)

Free space Dominating factor 187 196 205 210

downlink

Atmospheric Relatively small and 0.1 0.2 0.3 0.5

nearly constant at

high elevation angles

Rain attenuation Severe as frequency 0.1 0.5 2 6

and elevation angle

increases

Refraction Significant at times at 6 3 2 1

low elevation angles

Scattering Produces local RFI

Diffraction Considered for 6 to 12

mobile links without

line of sight

Ionospheric Occasional wide 3 to 6 1 to 3 <1

scintillation signal variation

(multipath)

Doppler Frequency shift for

moving vehicles or

satellites

The free-space path loss, which would be in the denominator of the preceding

equation, is

a0 = (4_R/_ )2

where _ = the wavelength in meters and R = the path length in meters.

Substituting frequency for wavelength (e.g., f = c /_ ) and converting to decibels,

we obtain the following basic relationship for free space loss:

A0 = 183.5 + 20 log( f ) + 20 log(R/35,788)

where f is the frequency in gigahertz and R is the path length in kilometers. (The

multiplier of 20 for the log accounts for the squaring of the terms within the

brackets.)

This formula is plotted at three downlink frequencies as a function of path

length in Figure 4.14. The figure illustrates the basic free-space loss that must be

countered as a function of the particular orbital configuration. In particular, we

see a fundamental difference of 30 dB between GEO and LEO satellite systems.

4.2.5.2 Absorption

RF energy is absorbed to some extent (and converted into heat) as it passes through

clear air, water vapor, and smog. The impact of frequency of operation is shown

in Figure 4.15 for an Earth station at sea level and transmitting or receiving at an

126 Microwave Link Engineering

Figure 4.14 Free-space loss, A0, as a function of orbit altitude and frequency band.

Figure 4.15 Typical relationship between atmospheric absorption and the microwave frequency

of transmission, expressed in decibels.

elevation angle toward the zenith (_ = 90°). The absorption essentially is constant

and will be under 1 dB at C- and Ku-bands. The only place on the ground where

_ = 90° is at the subsatellite point, on the equator directly below the satellite. At

other locations the elevation angle takes values down to perhaps a practical low

of 5°. At a particular frequency shown along the X-axis, the atmospheric absorption

for elevation angles less than 90° would increase due to the greater thickness of

atmosphere. Below 10°, absorption is highly sensitive to multipath due to ground

reflections and atmospheric ducting. The following formula indicates how atmospheric

absorption changes for elevation angles greater than 10°:

Aa

8aO + 2aW

sin _

4.2 Propagation on the Earth-Space Link 127

The two parameters, aO and aW, represent constituent absorptions due to

oxygen and water vapor, respectively. We can view the numerator as being the

total attenuation indicated on the Y-axis of Figure 4.15.

As frequency increases above 15 GHz, the constituents of the atmosphere reach

individual points of resonance, and absorption can become very high, even total.

The bands of fequencies around 22 and 66 GHz correspond to resonances for

water vapor and oxygen, respectively, and are not employed for either uplinks

or downlinks. Direct links between satellites, called intersatellite links (ISLs) or,

alternatively, cross-links, bypass the atmosphere and hence may utilize the absorptive

bands.

4.2.5.3 Rain Attenuation

After free-space loss, the most detrimental effect on commercial satellite links

above C-band is rain attenuation, which results from absorption and scattering of

microwave energy by rain drops. That loss, which increases with frequency, was

discussed in Chapter 1 in the comparison of frequency bands. Rain attenuation is

not predictable on an instantaneous basis, but statistical estimates can be made

that allow links to be designed. Obviously, dry seasons and regions of the world

with low rainfall would not suffer greatly from this phenomenon. However, links in

regions with heavy thunderstorm activity—and hence rainfall—should be provided

with greater link margin, or service might not be maintained with sufficient availability

to satisfy commercial requirements.

Intense rain is contained in rain cells, which have somewhat limited geographic

size. The statistical relationship between local rainfall and the resulting attenuation

caused at a particular microwave frequency is a complex combination of several

factors. The dimensions of the rain cell vary based on the rain rate, measured

typically in millimeters per hour. At a particular rain rate and cell size, the attenuation

increases with path length through the cell. Because a cell is shaped like an

oblate spheroid (i.e., flat like a hamburger), the path length is inversely related to

the elevation angle. Thus, rain attenuation varies in much the same manner as does

atmospheric attention in the absence of rain. Other factors include gaseous water

vapor, clouds, and ice particles. The work of combining all of these factors is

largely done and available to us in the form of rain propagation models. The

ITU has adopted the DAH model, which was developed by the research team of

Dissanayake, Allnut, and Haidara. Figure 4.16 provides in summary form some

rain attenuation data as a function of frequency and for a temperate climate such

as that of Western Europe or the northeastern United States. Comparable data for

a tropical region would be two to three times greater in decibels. The best approach

is to use a software tool like SatMaster (discussed later in this chapter) that uses

the DAH model to calculate rain attenuation for a specific location.

As indicated in the figure, the lower the elevation angle from the ground, the

greater is the amount of attenuation from a given amount of rainfall. Heavy rainfall

also alters the polarization of the signal because atmospheric drag causes raindrops

to flatten and not be perfectly spherical. That reduces cross-polarization isolation

between linear polarized transmissions. For example, the maximum isolation in

Figure 4.11 decreases at Ku-band in heavy rain from 40 dB to approximately

128 Microwave Link Engineering

Figure 4.16 Rain attenuation versus frequency and elevation angle for a temperate climate.

25 dB. That would not be detrimental to most transmissions if appropriate adjustments

are made in the link design. Depolarization in rain is not particularly harmful

at C-band and the lower parts of Ku-band, but it can greatly reduce isolation at

frequencies above 15 GHz. It also turns out that depolarization of CP transmissions

is somewhat more severe than for LP.

The first step in using a rain model is to identify the particular rain region

where the service must be provided. The map in Figure 4.17 shows different regions,

each characterized by a type of climate and associated rain properties. Tropical

regions are the worst, due to a higher incidence of thunderstorms, which contain

the most intense rain cells. The shading and numbers designate the relative impor-

Figure 4.17 Regions corresponding to the five rain-climatic zones, indicating the rain rate in

mm/hour to be exceeded for 0.01% of the time [2].

4.2 Propagation on the Earth-Space Link 129

tance of rain in the service design. This presentation is not detailed enough to

indicate higher rain rates in the most intense areas of Brazil, Indonesia, and central

Africa, where they reach 145 mm/hr for 0.01% of the time. Note that the attenuation

is approximately proportional to rain rate.

4.2.5.4 Refraction

The lower portion of the atmosphere, called the troposphere, decreases in density

upward from the Earth’s surface. As illustrated in Figure 4.18, electromagnetic

waves are bent as they pass through the medium; as a result, the satellite has a

virtual position slightly above that of its true position. That characteristic is taken

into account in terrestrial microwave system design by increasing the Earth’s radius

by a factor of 4/3 in the profile map used to plot the line-of-sight path. In satellite

links, the bending is less significant because of the higher angle toward the satellite.

Unstable atmospheric conditions like temperature inversions, clouds, and fog

produce discontinuities and fluctuations in what otherwise would be a uniform

air density distribution. The consequent random, temporary bending can cause

significant signal fading akin to rain attenuation. The effect is more pronounced

for paths that are nearly parallel to the Earth (line-of-sight microwave and satellite

links at low elevation angles). That is also a worry for making precise gain measurements

of an Earth station antenna when the reference transmitting source is on a

ground-based tower. Generally, refraction by itself does not impair typical satellite

links because the amount of bending is small relative to the beamwidth of the

satellite and the Earth station antennas.

The ionosphere, the layer of ionized particles at around 150 km, completely

reflects frequencies between 0.1 and 30 MHz under certain conditions, producing

the ‘‘sky wave’’ effect, which allows short-wave transmissions to cover long dis-

Figure 4.18 Refraction is the bending of a microwave signal as it passes through the layers of the

atmosphere.

130 Microwave Link Engineering

tances. In addition to refraction, the ionosphere rotates the linear polarization of

microwave transmissions (Faraday effect), although that can be compensated for

at frequencies above 3 GHz by appropriate adjustment of feed-horn angle at the

receiving Earth station. This is complicated by the fact that the angle of rotation

varies from day to night by as much as 6°. Below 3 GHz, Faraday effect produces

one or more full rotations, which is why CP is preferred. Faraday rotation increases

during periods of high sunspot activity, according to the 11- and 22-year solar

cycles. Faraday effect decreases with frequency and can be ignored at Ku-band and

above.

4.2.5.5 Scattering, Diffraction, and Shadowing

It was mentioned previously that water droplets scatter microwave signals. That

would reduce the direct-path power level, spraying some of it back toward the

source. At an Earth station site, the occasional scatter can place RFI in the direction

of terrestrial microwave receivers that otherwise would have been adequately

protected.

Diffraction, on the other hand, occurs when microwaves encounter and bend

over a physical obstacle such as a building or mountain. The principle behind

diffraction is illustrated in Figure 4.19, which shows how an Earth station might

be shielded from local microwaves by an intervening hill or mountain. The heights,

H1 and H2, introduce shielding but also provide the possibly diffraction path for

microwave signals in both directions. If the hilltop just happens to touch the lineof-

sight path between the Earth station and local microwave antennas (e.g., H1

and H2 are zero), a diffraction loss of precisely 6 dB is introduced. As the obstacle

begins to block the path, the amount of diffraction loss increases until there essentially

is total blockage.

The line-of-sight nature of microwave transmission means that blockage may

reduce signal strength below the threshold level. For frequencies above 3 GHz,

such blockage must be assumed whenever there is an obstacle that results in a

situation like that illustrated in Figure 4.19. The situation at lower frequencies

such as L- and S-bands might be somewhat more favorable because of the significantly

larger wavelength involved. Depending on the type of ground antenna in

use, it might be possible to receive adequate signal strength from reflections off of

buildings and hills. It has been proposed that partially blocked and reflected signals

Figure 4.19 The use of terrain shielding to block RFI between a terrestrial microwave link and an

Earth station.

4.2 Propagation on the Earth-Space Link 131

be used to directly communicate or to augment communications for mobile users

either with handheld satellite phones or in vehicles. This is certainly done in terrestrial

cellular systems, but they have the advantage of much higher link margins

from closely spaced base stations. Using a GEO or LEO satellite, it is not feasible

to obtain such high margin and so one cannot expect reliable communications under

shadowing conditions. The one area where some shadowing could be tolerated is

where it is produced by trees that are relatively far apart. Even here, heavy tree

trunks will block signals much the same as a building.

4.2.5.6 Multipath

Multipath is the result of the same RF signal taking both a direct path and a slightly

longer refracted path, the latter arriving at the receiving antenna delayed in time

from the former. It is illustrated in Figure 4.20 for refraction caused by discontinuities

in the ionosphere. Multipath also occurs under shadowing conditions. It will

be shown that both signal cancellation and enhancement can occur. The twinkling

of a star is a multipath phenomenon, where two light rays, one direct and the

other bent back in the troposphere, reach the eye and combine to produce a

variation in light intensity. The twinkling occurs when the variability in the air

causes the refracted path to change over time.

Due to the fact that rays are wave phenomena, the direct and refracted paths

combine using vector addition, as illustrated in Figure 4.21. Shown at the right

are the two signal paths reaching the same antenna, wherein they combine to form

a resultant signal (that is because both paths contain the same signal, although the

refracted path, being longer, introduces a delay and consequent phase shift). The

two drawings at the left of Figure 4.21 represent the extremes: enhancement at the

top and cancellation at the bottom. The direct path is shown as the vertical signal

vector of constant length. Added to that on a vector basis is the refracted signal

(the dotted vector), which can arrive at any random relative phase angle, as indicated

by the circle about the tip of the direct path vector. The resultant received signal

vector, shown as a heavy arrow, is only slightly affected by the weak refracted

Figure 4.20 Multipath propagation caused by ionospheric scintillation.

132 Microwave Link Engineering

Figure 4.21 Multipath signals produce enhancement or cancellation of the direct path signal when

received through a common antenna, as illustrated by vector diagrams.

signal. An increase or a decrease in strength occurs, depending on the relative phase

angles.

If the refracted signal is comparable in strength to the direct path, cancellation

occurs when the relative phase is 180 degrees. There also can be significant amplification

(up to 6 dB) when the two vectors instantaneously combine in phase.

Fortunately, the unstable conditions that produce this in the ionosphere only occur

during two periods annually, around the equinoxes. Earth stations near the geomagnetic

equator experience the most severe scintillation fades during those periods

of activity. Ionospheric scintillation is not constant, with the result that the received

signal both decreases and increases rapidly. The fact that the combining is frequency

dependent produces selective fading, in which a spectrum of signals fluctuates

widely. That is in contrast to shadowing due to terrain blockage, termed flat fading

because it is not frequency selective.

Finally, multipath is an influence on L- and S-band mobile links because the

user antenna is generally not very directional so can receive signals from several

directions at the same time. Generally, the direct path dominates, but the reflected

path or paths can produce pronounced fading. Terrestrial mobile radio services

do not depend on the line-of-sight path and actually utilize multipath to provide

coverage in areas of heavy terrain blockage. There, as much as 30 dB of margin

is afforded by the close range between mobile user and lease station.

Satellite links, on the other hand, do not normally generate 30 dB of margin

and so cannot depend heavily on multipath as a positive factor. Rather, signals

reflected off buildings and local terrain features more likely will cause fading. The

basic problem is illustrated in Figure 4.22 for a mobile receiver traveling along a

highway. Two paths reach the mobile user: the direct line-of-sight path and a path

that is reflected off the highway itself. They produce either fading or enhancement,

4.3 Microwave Transmitters and Receivers 133

Figure 4.22 Generation of mobile multipath on flat, smooth terrain.

which is nearly constant as long as the satellite remains in the same position relative

to the vehicle. The amount of loss depends on the reflection coefficient experienced

by the reflected signal. Alternatively, reflection off a fixed feature, such as a hill

or a building, produces time-varying fading as the vehicle moves relative to the

reflection point. The rapidity of the fading depends on the speed of the vehicle.

The design of MSS links is, therefore, the most challenging, because they are subject

to greater variation and uncertainty.

4.3 Microwave Transmitters and Receivers

The basic elements in an end-to-end satellite communication link are illustrated in

Figure 4.23. In the figure are the transmitting Earth station, which establishes the

uplink path, a simplified satellite and its microwave repeater, the downlink path,

and a receiving Earth station.

The entry and exit points to the propagation medium are provided by the

transmitting and receiving antennas. As previously discussed, antennas convert

electrical energy at microwave frequencies into electromagnetic waves, and vice

versa. A transmitting Earth station consists of equipment that impresses (modulates)

the information to be sent on an RF signal called the carrier, translates it to the

appropriate frequency and amplifies it to a high-enough power level to provide an

adequate uplink. A receiving Earth station works in the exact opposite direction.

Deferred to Chapters 6 and 9 are detailed descriptions of the electronics of the

satellite and the Earth station, respectively.

4.3.1 Transmitting Station

Figure 4.24 portrays a single transmitting chain of a generic microwave station,

with the signal input at the left and the RF output from the transmitting antenna

at the right. The signal to be transmitted consists of information in electrical form,

such as one or more voice channels for telephone service, digital data in the form

134 Microwave Link Engineering

Figure 4.23 The main elements of a satellite link on an end-to-end basis.

4.3 Microwave Transmitters and Receivers 135

Figure 4.24 Simplified block diagram of a microwave transmitting station capable of baseband

signal processing, modulation, frequency translation, and high power amplification.

of a high-speed bit stream, or a composite video signal such as that delivered from

a video tape recorder. In modern satellite systems, analog information forms like

voice and TV are first digitized and then compressed to reduce the required bandwidth.

From that point, the link is digital in nature.

4.3.1.1 Encoding and Modulation

In the encoding element of the station (the first element to be encountered), the

digital information is prepared for transmission through the link. Encoding consists

of one or more numerical processes that better match the data to the specific

characteristics of the satellite link. Table 4.4 summarizes the most common forms

Table 4.4 Typical Forms of Encoding Applied to Microwave Links over Communications

Satellites

Impact on Data

Encoding Technique Application Benefits Throughput

Forward error Error rate reduction Lower error rate, or Increases the output

correction reduced power bit rate to

requirement (Eb /No ) accommodate

redundant bits

Compression Reduces the total bit Less data to send or Increases throughput;

count, either lossless store—better may reduce quality

or lossy utilization of the link and introduce delay

and system

Encryption Information security Makes data private Generally, no increase

and difficult to in data rate; involves

corrupt complex management

and could introduce

delay

Protocol adaptation Data communications Improves the user May increase actual

networks, typically experience by throughput;

using the Internet countering satellite introduces

Protocol suite link impairments, complexity; tied to the

(TCP/IP) such as bit errors and type of information

delay transfer or application

136 Microwave Link Engineering

of encoding along with their benefits and characteristics. Forward error correction

(FEC) is one type of encoding that is always applied as it can only improve the

quality of transmission in terms of the bit error rate (BER). A reduction in BER

can be taken either as a quality enhancement or to reduce the amount of transmitted

power (alternatively, the antenna size). Modern FEC techniques use combinations

of individual error correcting codes through the process of concatenation.

In many cases, FEC is incorporated into the modulation and demodulation

function, which are performed by the modem. Compression techniques fall into

two categories: lossless compression, which only removes useless bits that convey

no information, and lossy compression, where information is removed but is

intended to provide sufficient quality to satisfy the user and application. Information

that starts out in analog form, such as voice, image, and video, can employ lossy

compression as users are accustomed to some amount of impairment. However,

the amount of compression deserves careful review by the intended audience to

assure that the service is not compromised as to acceptability. Encryption is highly

desirable on satellite links as it is a relatively simple matter to intercept satellite

transmissions and potentially to introduce either bogus data or to otherwise disrupt

information transfer. Protocol adaptation was discussed in Chapter 3 and is very

effective for reducing the perceived delay of Internet services over GEO satellites.

The output of the encoding stage is referred to as the baseband. It is the purpose

of the modulator to take the baseband and apply it to an RF carrier. The type of

modulator used in a microwave station often handles a wideband baseband input

such as that obtained from a high-capacity data stream measured in megabits per

second. In SCPC service, the output of the modem is kept on to allow a continuous

stream of data to be uplinked to the satellite. TDMA operation, on the other hand,

requires that the modem transmit in noncontinuous bursts. That is because the RF

channel is being shared by multiple Earth stations that transmit in different time

slots.

The opposite of modulation is simply demodulation, which is the process

whereby the baseband is removed from the carrier. The demodulator intially must

acquire the incoming carrier, demodulate the bit stream, and then produce the

baseband in a form that can be used by the decoder. The process is repeated

for each received transmission, particularly for TDMA operation. The decoding

circuitry corrects a majority of the errors produced by noise and interference on

the uplink and downlink.

4.3.1.2 Frequency Conversion and RF Amplication

The RF carrier coming from the modulator typically is not at microwave frequencies

but rather is centered within a standard frequency channel, the intermediate frequency

(IF). Most transmitting and receiving stations use 70MHzas the IF, allowing

modulators and demodulators to be conveniently interchanged and interconnected

by patch cords and coaxial switches. In low-cost consumer equipment, the IF is

internal to the unit and hence need not follow any particular standard. Another

point is that there are cases where the RF bandwidth is larger than 140 MHz,

making it unfeasible to use 70 MHz as the IF (since the bandwidth would extend

below zero frequency). This can be overcome by directly modulating a carrier at

4.3 Microwave Transmitters and Receivers 137

the microwave frequency of transmission or, more likely, by using an adequately

high IF, such as 140 MHz.

For digital modulation, the bandwidth is directly proportional to the input

data rate:

B = am _ R

where R is the data rate in bits per second and am is a constant determined by the

particular type of digital modulation and FEC. For example, QPSK is a popular

form of digital modulation used on satellites, in which case am is nominally equal

to 0.6. The proportionality constant, am, also depends on the order of the modulation.

For example, going from QPSK to 16 QAM cuts am in half, to 0.3. Employing

FEC with a code rate of 3/4 increases am to 0.8 and 0.4 for QPSK and 16 QAM,

respectively.

The function of the upconverter is to translate the carrier without modification

from IF to the desired microwave frequency of transmission. Within the upconverter

are a microwave mixer and a local oscillator (LO). Translation is governed by a

simple mathematical relationship that states that the output frequency equals the

sum of the input IF and the frequency of the LO. For example, if the IF is at

70 MHz and the LO is at 6,030 MHz, the output RF is 6,100 MHz. (Although

in theory the difference between the LO and IF could be employed, this is not done

in practice due to more complex filtering.) From a practical standpoint, the RF

usually is assigned by someone else and the IF is fixed. Therefore, the LO frequency

must be selected properly to put the RF carrier in the right place (e.g., on the

assigned frequency, which is 6,100 MHz in the example). Modern upconverters

are frequency agile, which means that the LO can be tuned for a different RF

channel from the front panel in much the same way as a TV set or car radio. The

particular type of LO is called a frequency synthesizer because the frequencies are

generated digitally using computational techniques. A nonsynthesized type of agile

LO could cover the frequency range but would not be adequately stable over time

and temperature. The desired frequency would be selected either from the front

panel using an input keypad or remotely by computer over a data line.

The design of the IF aspect of Earth stations has evolved to a more flexible

concept using a wideband IF nominally within the L-band within the broad range

of 0.5 to 2.5 GHz. This is a continuous spectrum that is not directly radiated and

so does not interfere with MSS services within this range. Using low-cost coaxial

cable (coax), this L-band IF carries 500 MHz of bandwidth between the outdoor

antenna system and indoor electronics used to select the specific transponder and

frequency, and for carrier demodulation and baseband processing. The appropriate

receiving device at the antenna is called a low-noise block converter (LNB), which

contains a low-noise amplifier (discussed later in this chapter) and a block downconverter.

On the transmit side, there would need to be an upconverter and highpower

amplifier; if the transmit power required is less than about 10W, then it is

possible to obtain both functions within what is called a block-upconverter (BUC).

4.3.1.3 High-Power Amplification

The last active element of the transmitting station is the high-power amplifier

(HPA). Because all processing and frequency translation have been accomplished

138 Microwave Link Engineering

in prior stages, the only function of the HPA is to increase the power of the

microwave carrier from the low output of the upconverter to the power level needed

to achieve satisfactory uplink operation. The HPA must have sufficient bandwidth

to operate at the assigned microwave frequency and cover the active RF bandwidth

of any anticipated carrier type. Examples of typical microwave HPA devices and

their respective power capabilities are shown in Figure 4.25. Vacuum-tube HPAs,

which do not require water cooling, offer the highest power. They include the

klystron power amplifier (KPA) and the traveling wave tube amplifier (TWTA).

Low to moderate power is affected by solid state power amplifiers (SSPAs), which

are designed for a variety of power levels.

The KPA has a capability measured in kilowatts and is popular for video

uplinks. Within the klystron microwave tube there is a resonant waveguide cavity

that is tuned to the specfic frequency of operation. The operating bandwidth of a

KPA is in the range of 50 to 100 MHz, making it necessary to retune the internal

structure to change transponders. That difficulty is overcome with the TWTA,

another class of microwave amplifier that happens to be very common on satellites

as well (see Chapter 6). TWTAs have a higher-percentage bandwidth, which is the

ratio of usable bandwidth to the center operating frequency, than KPAs and so

can transmit over a total bandwidth of 500 MHz at C-band and as much as 2,000

MHz at Ku- and Ka-bands. Practical TWTAs can be found with power outputs

of from 50W to as much as 800W, although 10-kW water-cooled TWTAs were

used in early INTELSAT Earth stations. Being vacuum tubes, the TWTA and the

KPA require sophisticated high-voltage power supplies, and both employ heated

cathodes to emit electrons for use in the process of amplification. In the case of

HPAs used in Earth stations, it is unavoidable that both types of high-power tubes

will wear out and need to be replaced after a few years of operation. TWTAs used

in satellites must have longer lifetimes, lower mass, and greater dc-to-RF efficiencies

Figure 4.25 RF output capability at C-, Ku-, and Ka-bands of standard uplink microwave power

amplifiers.

4.3 Microwave Transmitters and Receivers 139

and so have to be designed differently. That also results in lower power outputs,

generally under 300W at the time of this writing.

In low-power applications, such as the VSATs mentioned above, SSPAs can

be employed for stable, long-term operation without maintenance. A detailed discussion

of SSPAs can be found in Chapter 6. Figure 4.25 presents SSPA power

levels available in 2008: up to 800W at C-band, 250W at Ku-band, and 20W at

Ka-band. The basic building block of the SSPA is the gallium arsenide field effect

transistor (GaAsFET), and the overall power capability is determined by the number

of individual transistor stages that can be effectively paralleled. Bandwidth of SSPAs

is somewhere between those of the KPA and the TWTA, based on the biasing and

tuning of the transistor stages. That generally cannot be altered after the SSPA is

manufactured.

4.3.2 Receiving Station

The reverse process found in the receiving station is illustrated in Figure 4.26.

Because the microwave signal collected by the receiving antenna is weak, it first

is necessary to raise the power to a level that can be accommodated by the processing

elements. That is performed by the low-noise amplifier (LNA), whose gain must

meet the requirements described above. However, the internal noise contribution

of the LNA must be held small enough, or the weak signal input literally can be

buried in noise. The rest of the elements perform functions that are inverse to those

of the transmitting station. If an integrated downconverter is appended to the LNA,

the resulting unit is called a low-noise block converter (LNB).

4.3.2.1 Low-Noise Amplifiers

It is possible to express noise in terms of an equivalent noise temperature in Kelvin

(K). The Kelvin scale has gradations the same size as degrees C (centigrade) but

begins at the noiseless state of absolute zero and measures the average random

energy of motion of electrons within the receiver electronics. According to the

theory, the random energy of the electrons is proportional to the noise power that

overlays the desired signals within the passband of the amplifier. A device cooled

to 0K (an impossibility) would not contribute noise to the system.

Figure 4.26 Simplified block diagram of a microwave receiving station capable of low-noise amplification,

frequency conversion, demodulation, and baseband signal processing.

140 Microwave Link Engineering

All LNAs and LNBs found in satellite communications Earth stations today

use low-noise transistors such as gallium arsenide field effect transistors (GaAs-

FETs). These are also referred to as high electron mobility transistors (HEMTs),

which are more efficient and produce less noise than their silicon bipolar transistor

counterparts (although silicon transistors are more rugged and static discharge

resistant). Most transistors used in large-scale integrated circuits are FETs as are

microwave amplifiers of various types. Indium phosphide is another material used

in HEMTs and is gaining in popularity due to its improved performance over

GaAsFETs. Typical values of LNA or LNB noise temperature at C-, Ku-, and

Ka-bands are 30K, 60K, and 90K, respectively. Another important property of the

LNA and LNB is its gain, which can be selected in the range of 50 to 70 dB. This

involves one or two low-noise HEMT stages followed by one or more mediumpower

amplification stages that can be either HEMT or silicon. An LNB also

incorporates the block downconverter stage that employs a wideband mixer and

a local oscillator. The latter has a significant impact on the performance of the

Earth station, particularly for reception of narrowband signals that cannot tolerate

frequency instability or the oscillator impairment called phase noise.

It is common practice in Earth station design to place the LNA as close as

possible to the feed and thereby minimize the input loss. That has two benefits:

(1) it improves the effective gain of the receiving system because any loss on the

input must be subtracted from that of the antenna, and (2) reducing this input loss

also decreases the effective noise of the receiving system. The latter effect is discussed

in later chapters.

4.3.2.2 IF-to-Baseband Chain

As discussed in Chapter 1, the satellite’s downlink frequency usually is offset from

its uplink frequency to prevent self-interference on the satellite (the exception is

for single-band systems like Iridium that sequentially transmit and receive packets

on the same frequency). To receive the carrier, the frequency of the LO of the

downconverter would differ from that of the upconverter by the fixed frequency

offset (2,225 MHz for standard C-band) introduced by the satellite repeater. To

complete a previous example, the downconverter LO should be set to the (6,100

2,225) the IF.

The demodulator is tuned to the IF to produce the baseband from the received

carrier and sometimes provides FEC decoding. Along with the information are

versions of noise and interference picked up over the link. Specifically, the received

digital carrier on the downlink is demodulated to deliver a digital bit stream with

the digitized version of the desired information. The demodulated performance is

measured in terms of bit error rate. The bits are processed further to reduce the

error rate, decompress the underlying information, and, if appropriate, convert the

data back into the original analog form. Greater detail on digital modulation

characteristics is contained in Chapter 5.

4.3.3 Definition of a Transponder

The satellite communication industry has long used the term transponder in reference

to a defined RF channel of communication within the communication payload.

4.3 Microwave Transmitters and Receivers 141

The term itself is a contraction of transmitter-responder, originally referring to a

single-frequency repeating device found on aircraft. The purpose of the aircraft

transponder is to add the identification of the aircraft and actively enhance the

power to be reflected back to the radar transmitter. A satellite transponder is

entirely different because it is more of a transparent microwave relay channel, also

taking into account the need to translate the frequency from the uplink range to

the downlink range.

We can better define a transponder by examining the two different payload

configurations shown in Figure 4.27. The single-channel repeater (shown in A)

does just what its name implies: provide a single channel of transmission within

the satellite. As shown at the top of the figure, the entire uplink band is translated

in the downconverter and applied to a single power amplifier. As a consequence, this

amplifier must accommodate the entire uplink bandwidth, which would amount to

500 MHz at C- or Ku-band. That tends to limit the power-handling capability,

particularly if multiple carriers are amplified by the common output stage. The

first generation of LEO and GEO satellites like Telstar and Syncom, respectively,

had repeaters of this type, capable of carrying one TV channel each.

The transponderized design (shown in B in Figure 4.27) breaks up the downlink

range into individual frequency channels. Figure 4.28 suggests how eight transponders

would be divided from the uplink bandwidth, indicating the usable transponder

bandwidth and a guard band. Note that the uplink section still carries the entire

bandwidth of 500 MHz and is shared by all transponders. After the downconverter,

there is a bank of RF filters, called the input multiplexer (IMUX), that separates

the transponder channels on a frequency basis. The output of each IMUX filter

contains one transponder bandwidth, to be amplified in an individual power amplifier

(PA), either a TWTA or SSPA, depending on the frequency and power required.

On the output side, the RF power of the PAs is summed in a passive power combiner

called the output multiplexer (OMUX); it is composed of specially designed low-

Figure 4.27 General arrangement of (a) a single-channel repeater and (b) a channelized repeater

with multiple transponders.

142 Microwave Link Engineering

Figure 4.28 Hypothetical frequency plan illustrated for eight transponders in one polarization.

loss waveguide filters. A single waveguide transfers all the power from the OMUX

to the antenna system.

The transponder, then, is a combination of elements within the payload. On

the input side, it represents a share of the common uplink and receive equipment

within the repeater. We are able to identify specific equipment for each transponder

on the downlink side, consisting of the input filter, power amplifier, and output

filter. Not shown in the figure are the necessary spare active elements (redundancy)

to ensure continuity of service in the event of amplifier or receiver failure. Also

not shown at the input to each PA is a level controlling channel amplifier or

linearizer often found on modern satellites.

The fact that a transponder can be assigned to a particular user application

network has caused them to be rented or sold like condominium flats. In actuality,

it is the microwave channel of communication bandwidth that the transponder

lessor or purchaser acquires.

4.4 Overall Link Quality

One of the more complex problems in microwave link engineering is knowing

precisely how a particular signal will be affected by the noise, which is random in

nature, and by interference from other radio carriers. This section reviews some

of the factors that determine the actual quality and threshold for digital receivers.

A clear understanding of the threshold of the link allows the engineer to predict

how much margin will be available to overcome fading from the various sources

discussed at the beginning of this chapter. It is presumed that the circuitry involved

is properly impedanced matched, so that signal reflections do not unduly distort

the signals.

4.4.1 How Noise and Interference Affect a Microwave Link

Both noise and interference will degrade service quality and if not controlled will

at times render the link unusable. Interference, as explained previously, is often

4.4 Overall Link Quality 143

due to the RF transmission of someone else. On the other hand, noise is totally

random in nature because it results from the random motion of electrons or other

elemental particles in the environment or the receiving equipment. Pure white noise

is most common on satellite links and produces random voltage fluctuations whose

probability follows a normal (Gaussian) distribution. The term white refers to the

fact that the frequency spectrum does not have discrete components at specific

frequencies but rather is a continuum of frequencies, like white light. Ideal white

noise, running from zero to infinite frequency, is physically impossible since it

would represent infinite power. Noise is sufficiently white if it is constant over the

bandwidth of the signal in question.

The resulting density of the noise power (noise power density) in watts per

hertz is proportional to the equivalent noise temperature in Kelvins, that is,

N0 = k _ T

where k is Boltzmann’s constant (i.e., 1.3803 _ 1023 W-sec/K or 228.6 dBW-sec/K

when expressed in decibels).

This simple relationship demonstrates why the noise performance of an LNA

of a receiving Earth station is rated in terms of an equivalent noise temperature.

The actual noise power that affects a given signal is that which lies with the signal’s

bandwidth, B, that is,

N = k _ T _ B

The effect of white noise on a modulated carrier containing binary data can

be shown using time waveforms, presented in Figure 4.29. The noise-free case is

shown in Figure 4.29(a) in the form of digital information impressed on a carrier.

The transition from binary 1 to 0 occurs where the phase of the sinewave reverses.

That type of modulation is called phase shift keying (PSK) because the shifting

(flipping) of phase by 180 degrees is the means by which information is transmitted.

It is the job of the demodulator to detect the phase reversal to convey to the user

Figure 4.29 (a–c) A digital phase-shift keyed signal combines with white noise to produce a

distorted signal waveform that must be detected by the receiver.

144 Microwave Link Engineering

the proper bit sense. The PSK signal is sent through the link and enters the receiver

along with white noise inherent in the electronics. That noise is depicted in Figure

4.29(b), where the mean voltage is zero and the standard deviation of the noise is

approximately one-half the carrier amplitude. Because power is proportional to

the square of the voltage, the true ratio of carrier power to noise power is 4, or

equivalently 6 dB. The rather high relative noise (low C/N) level yields the sum of

carrier plus noise shown in Figure 4.29(c), where the carrier appears to be somewhat

obliterated by noise spikes.

A well-designed demodulator can detect the transition from the 1 to 0 state in

most cases. The noise can occasionally reach a voltage greater than the signal

amplitude; if it has the opposite sense to the transmitted signal at the precise

moment of testing, it can cause the demodulator to make the wrong choice. The

assumed binary digit will then be incorrect (i.e., a 1 instead of a 0 or vice versa,

depending on what was sent). It should now be possible to visualize that the

strength of the signal relative to the noise will determine the rate at which errors

in detection are made. This is a significant area of engineering study in the communications

field. Some of the most important applications of digital signal processing

(DSP) allow low-cost sending and receiving equipment to identify errors and reverse

them prior to delivery to the user. Special codes that provide such forward error

correction (FEC) accomplish this in practice.

Wideband interfering signals can be treated as white noise and often are.

However, if the interference is narrower in bandwidth than the desired signal, more

sophisticated analysis techniques or even direct measurement must be employed.

As that type of investigation is beyond the scope of this book, we will assume that

the interference can be converted into an equivalent amount of white noise and

added on a power basis to the other noise within the bandwidth of the desired

signal.

4.4.2 Carrier-to-Noise Ratio

The strength of the carrier relative to the noise and not the absolute level of what

comes out is what determines the quality of transmission. This can be understood

by realizing that the power level of the baseband signal from the receiving station’s

signal processor can be raised by simple power amplification. However, the noise

that is present also will be amplified. Only by suppressing the noise at the input

to the receiver can the quality of transmission be improved. That is why the true

performance of the link is measured by the ratio of RF carrier power to noise

power. A variant used in digital transmission is the ratio of the energy per bit to

the noise power density, Eb /N0 . It will be shown that there is a simple constant

adjustment factor between C/N and Eb /N0 , that is,

C

N =

Eb

N0

_

Rb

B

where Rb is the information bit rate, and B is the carrier bandwidth.

A receiving system showing a carrier, white noise, and a single source of

interference is presented in Figure 4.30. The frequency spectrum is similar to the

4.4 Overall Link Quality 145

Figure 4.30 Frequency spectrum showing the desired carrier and the interfering carrier and white

noise, as seen on a spectrum analyzer.

display on a microwave spectrum analyzer, a useful piece of test equipment. Spectrum

analysis is vital to microwave communications because it provides one of the

few windows for viewing the reception process. The figure shows a constant spectrum

of white noise, providing what is referred to as a noise floor. The vertical

scale is linear, measuring power density in terms of watts per hertz. Piercing through

the floor are the desired carrier occupying bandwidth B and centered at frequency

F2, along with an interfering carrier at frequency F1. The total noise power, N,

over the bandwidth of the carrier is the product N0B. An actual spectrum analyzer

would have a vertical scale measured in decibels to simplify the measurement of

C/N. In that case, the C/N in decibels is equal to the difference between the

measured decibel levels of total carrier power and noise power within the bandwidth

B. While bandwidth can be taken into account by subtracting the quantity

10 log B from the spectrum analyzer reading, it is also possible to present the

spectrum analyzer’s measurement bandwidth to eliminate the need for manual

correction.

Another correction to the measurement of C/N on a spectrum analyzer results

from the fact that this instrument actually displays the sum of carrier power and

noise power, since the noise is present within the carrier bandwidth. This correction

has the following form:

(C + N)

N =

C

N +

N

N

=

C

N + 1

C

N =

(C + N)

N 1

146 Microwave Link Engineering

This formula is performed on true ratios of powers, and not in decibels. To

perform this calculation, take the reading in decibels, convert to a ratio, subtract

1, and then convert back to decibels. For example, if (C + N)/N equals 8 dB (a

factor of 6.3), then the actual C/N equals 5.3 as a ratio or 7.2 dB.

The type of interference shown in Figure 4.30 is not noiselike and therefore

cannot be analyzed in a straightforward manner. One consideration is that only

the upper corner of the interference actually affects the carrier. The ratio of carrier

power to interference power can be taken from the spectrum analyzer display, but

its effect on the reception process also depends on the frequency offset (F2 F1)

between the two carriers and their relative bandwidths. If due to a malfunction,

the interference may rove through the bandwidth for the carrier, affecting it as

well as other carriers on adjacent frequencies.

4.4.3 Link Budget Analysis

The key station elements and propagation phenomena having been described, it is

now possible to review the analysis and prediction of link performance. Figure

4.31 shows a simplified microwave link with a key parameter (in dB) indicated

for each element. The transmitter can be characterized by the HPA power output,

with the transmitting waveguide introducing some loss as it carries the power to

the transmitting antenna. The electromagnetic wave propagates outward from the

antenna into the medium, where it is subjected to various losses, the free-space

loss being dominant. The small amount of signal power gathered by the receive

antenna is carried through waveguide to the LNA of the receiver. It is possible to

characterize the receiver by the minimum acceptable threshold, which takes into

account the RF noise as well as the quality desired by the user.

4.4.3.1 Power Balance Equation

The simple mathematical relationship shown at the bottom of Figure 4.31 is the

power balance equation. The parameters of the link are actually factors: however,

it is a common practice to express all of them in decibels because that reduces the

Figure 4.31 The relationship between the critical elements of a microwave link and the power

balance equation.

4.4 Overall Link Quality 147

analysis to addition and subtraction. The power balance in decibels can be stated

simply: The power received equals the power transmitted plus all gains and minus

all losses. Transmitter power is expressed in decibels relative to 1W (dBW). For

example, 1W is by definition 0 dBW; 2W is 3 dBW; 10W is 10 dBW; 100W is

20 dBW; and so on. To preserve proper dimensions, there can be only one dBW

term on each side of the equation (the rest are dimensionless dB terms). The power

balance equation usually is arranged in a table called a link budget, in which each

paramter of the link is provided with its own explanation and quantity. One

approach to the problem is to set up for each case to be analyzed, using a spreadsheet

program like Microsoft Excel. There are specialized link budget programs on the

market such as SatMaster that simplify and standardize the process [3].

4.4.3.2 Typical Link Budget (Ku-Band)

A typical link budget for a Ku-band satellite downlink is presented in Table 4.4.

The first entry is the RF power output of the transmitter, expressed first in watts

and then converted into dBW. Transmit waveguide loss of 1 dB (26% power

reduction) is subtracted, while the gain of the spacecraft antenna in the direction

of the receiving Earth station is added. A detailed discussion of spacecraft antenna

design and performance is given in Chapter 7. It is customary to show a subtotal

at this point called the effective isotropic radiated power (EIRP), indicating how

the satellite is performing compared to an isotropic source with 1W of RF drive

power (i.e., 0 dBW). The value of 53 dBW shown in the table is typical for current

high-power Ku-band satellites in the BSS. For a large country such as the United

States, China, Indonesia, or Australia, this can be delivered by a 250W TWT.

With the satellite in GEO at an altitude of 36,000 km, the link budget contains

a single entry of 205.6 dB of free-space loss. In contrast to spreading loss (e.g.,

4_R2 ), path loss includes _ as a component, having the arithmetic form:

A = (4_R2)(4_ /_2 )

= (4_R)2/_2

converting to dB, A = 92.5 + 20 log F + 20 log R where F is the frequency in GHz

and R is the range in km.

Other propagation losses, such as rain attenuation, that are random in nature

are evaluated separately and compared against the overall link margin. Since this

is a clear sky link budget, we include an allocation of 0.1 dB for atmospheric loss.

The next two items relate to the receiving Earth station: the peak gain of an assumed

45-cm antenna and 0.5 dB of waveguide loss. The combined power balance yields

a received power of 120.5 dBW at the input to the Earth station LNA and receive

electronics. For that hypothetical link, assume we have a receive system noise

temperature of 140K and a signal bandwidth of 27 MHz. For the corresponding

received noise power of 132.8 dBW (e.g., kTB), the downlink C/N result is

12.3 dB.

Uplink noise within the spacecraft receiver contributes a lesser amount to the

total link noise and is ignored in the present example. However, the standard

148 Microwave Link Engineering

approach is to prepare a separate link budget for the uplink, following approximately

the same format as Table 4.5. Then the overall link C/N is determined

using the following formula:

C/Nt = (Nd /C + Nu /C)1

calculated as ratios and not decibels. The final step is to convert back to decibels.

Above, the subscripts t, d and u indicate total, down- and up-link components,

respectively.

The next three lines allow us to translate the expected clear-sky C/N into the

corresponding value of Eb /No . The process for doing this is to first take the ratio

of the carrier bit rate, here assumed to be 30 Mbps, and the carrier bandwidth.

That turns out to be 1.111, which corresponds to 0.5 dB of adjustment. This

is subtracted from C/N to obtain the Eb /No for the downlink under clear sky

conditions.

4.4.4 Link Margin

The final and perhaps key concept in microwave link engineering is that of link

margin, which is nothing more than the excess power in the carrier relative to the

threshold value. It is possible to measure the threshold in the laboratory under

controlled conditions. First, the link performance without fading can be determined

using the power balance equation and spectrum analysis. The margin, then, is the

difference in decibels between the minimum value and the threshold value. For

example, if the minimum acceptable (threshold) Eb /N0 is 8 dB, the link margin is

11.8 8 = 3.8 dB. After the link is operational, the margin can be verified by

manually decreasing the transmit power to the point where the received baseband

Table 4.5 Typical Ku-Band Link Budget for the Downlink to a Small DTH Receiver

Relative Value

(Expressed

Parameter as a Gain) Absolute Value Units

Transmit power (250W) 24.0 dBW

Transmit waveguide loss 1.0 dB

Transmit antenna gain 30.0 dBi

EIRP 53.0 dBW

Free-space loss 205.6 dB

Atmospheric loss 0.1 dB

Receive antenna gain (45-cm diameter) 32.7 dBi

Receive waveguide loss 0.5 dB

Received power (clear sky) 120.5 dBW

Bandwidth (27 MHz) 74.3 dB(Hz)

System noise temperature (140K) 21.5 dB(K)

Boltzmann’s constant 228.6 dB(Hz*K)1

Received noise power 132.8 dBW

C/Nd 12.3 dB

Bit rate (30.0 Mbps)

Ratio of bit rate to bandwidth (1.111) 0.5 dB

Eb /No (downlink) 11.8 dB

4.4 Overall Link Quality 149

signal is barely acceptable. The recorded change in transmit power in decibels is

essentially the link margin.

Adequate link margin allows the link to deliver exceptionally good quality

(low bit error rate) under mean signal, clear weather conditions. However, when

either the uplink or the downlink is experiencing heavy rain or mulipath fading,

the available link margin determines how often and for how long the link will

drop below threshold. The criterion often used is the availability, that is, the

percentage of time the link is above threshold. Typical satellite links operate in the

range of 99% to 99.95%, demonstrating the high reliability of line-of-sight paths

between the satellite and its associated Earth stations.

References

[1] Final Acts of WRC-77, 2. Radio Propagation Factors, International Telecommunication

Union, Geneva, 1997.

[2] ITU Radio Regulations, Vol. 2, Appendix S7, Edition of 2008.

[3] Sat Master software, available fromArrowe Technical Services (http://www.satmaster.com/).

 

  
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